Recent decades have seen a surge of interest in miniaturised electro-mechanical systems, so-called MEMs technology. One promising area of MEMs is that of microfluidics; the control, manipulation and sensing of micro-liter volumes of liquids. Applications in this area are numerous and include chemical synthesis and the chemical and biological analysis of small quantities of substances, so called Lab-on-a-chip (LoaC). A general introduction to the field can be found in many standard textbooks, e.g. “Introduction to Microfluidics”, Patrick Tabeling, Oxford University Press (Jan. 2006), ISBN 0-19-856864-9.
Digital microfluidics is concerned with the control, manipulation and sensing of droplets of fluids on an individual basis. An introduction to this area can be found in chapter 2 of “Microfluidics for Biotechnology”, Berthier and Silberzan, Artech House (2006), ISBN-10:1-58053-961-0.
This reference also describes the method of electrowetting on dielectric (EWOD) for the control and manipulation of droplets in an array-based architecture. A simple architecture for performing EWOD is shown in FIG. 1. A substrate 25 has disposed upon it a conductive electrode 22, with an insulator layer 20 deposited on top of that. The insulator layer 20 separates the conductive electrode 22 from the hydrophobic layer 16 upon which an ionic droplet 4 sits. The droplet 4 makes a contact angle θ with the surface of the hydrophobic layer 16, the value of which depends on the hydrophobicity of the surface. By applying a voltage V to the conductive electrode 22, the contact angle 6 (θ) can be adjusted. An advantage of manipulating contact angle by means of EWOD is that power consumption is low since there is no path for DC current to flow.
FIG. 2 shows an alternative and improved arrangement whereby a top substrate 36 is also supplied, containing an electrode 28 coated with a hydrophobic layer 26. A voltage V2 may be applied to the top electrode such that the electric field at the interfaces of the droplet 4 and hydrophobic layers 16,26 is a function of the difference in potential between V2 and V. A spacer 32 may be used to fix the height of the channel volume within which the droplet 4 is constrained. In some implementations the channel volume around the droplet 4 may be filled by a non-ionic liquid, e.g. oil 34. The arrangement of FIG. 2 is advantageous compared to that of FIG. 1 for two reasons: firstly it is possible to generate larger and better controlled electric fields at the surfaces where the droplet 4 contacts the hydrophobic layers 16,26. Secondly the droplet 4 is sealed within the device preventing losses due to evaporation etc.
U.S. Pat. No. 6,565,727 (A. Shenderov; issued May 20, 2003) discloses a passive matrix EWOD device for moving droplets through an array. The device is constructed as shown in FIG. 3. The conductive layer of the lower substrate 25 is patterned so that a plurality of electrodes 22, 38 are realised. These may be termed the EW drive elements. By applying different voltages, termed the EW drive voltages, (e.g. V and V3) to different electrodes (e.g. electrode 22 and electrode 38), the hydrophobicity of the surface of the hydrophobic layers 16,26 can be controlled, thus enabling a droplet 4 to move in the lateral direction. U.S. Pat. No. 6,911,132 (V. Pamula et al.; issued Jun. 28, 2005) discloses an arrangement, shown FIG. 4, whereby the conductive layer on the lower substrate is patterned to form a two dimensional array 42. By the application of time dependent voltage pulses to some or all of the different electrodes it is thus possible to move a droplet 4 though the array on a path 44 that is determined by the sequence of the voltage pulses. U.S. Pat. No. 6,565,727 further discloses methods for other droplet operations including the splitting and merging of droplets and this mixing together of droplets of different materials. In general the voltages required to perform typical droplet operations are relatively high. Values in the range 20-60V are quoted in prior art (e.g. U.S. Pat. No. 7,329,545 (V. Pamula et al., issued Feb. 12, 2008)) depending on the technology used to create the insulator and hydrophobic layers.
U.S. Pat. No. 7,255,780 (A. Shenderov; issued Aug. 14, 2007) similarly discloses a passive matrix EWOD device used for carrying out a chemical or biochemical reaction by combining droplets of different chemical constituents.
Thin film electronics based on thin film transistors (TFTs) is a very well known technology which can be used, for example, in controlling Liquid Crystal (LC) displays. TFTs can be used to switch and hold a voltage onto a node using the standard display pixel circuit shown in FIG. 5. The pixel circuit consists of a switch TFT 68, and a storage capacitor 58. By application of voltage pulses to the source line 62 and gate line 64, a voltage Vwrite 66 can be programmed and stored in the pixel. By applying a different voltage to the counter-substrate CP 70, a voltage is thus maintained across the liquid crystal 56 (represented by capacitance CLC) in the pixel location.
Many modern displays use an Active Matrix (AM) arrangement whereby a switch transistor is provided in each pixel of the display. Such displays often also incorporate integrated driver circuits to supply voltage pulses to the row and column lines (and thus program voltages to the pixels in an array). These are realised in thin film electronics and integrated onto the TFT substrate. Circuit designs for integrated display driver circuits are very well known. Further details on TFTs, display driver circuits and LC displays can be found in standard textbook, for example “Introduction to Flat Panel Displays”, (Wiley Series in Display Technology, WileyBlackwell (January 2009), ISBN 0470516933).
U.S. Pat. No. 7,163,612 (J. Sterling et al.; issued Jan. 16, 2007) describes how TFT based electronics may be used to control the addressing of voltage pulses to an EWOD array by using circuit arrangements very similar to those employed in AM display technologies. FIG. 6 shows the approach taken. In contrast with the EWOD device shown in FIG. 3, the bottom substrate 25 is replaced by a substrate 72 with thin film electronics 74 disposed upon it. The thin film electronics 74 are used to selectively program voltages to the patterned conductive layer electrodes 22,38 used for controlling electrowetting. It is apparent that the thin film electronics 74 can be realised by a number of well known processing technologies, for example silicon-on-insulator (SOD, amorphous silicon on glass or low temperature polycrystalline silicon (LTPS) on glass.
Such an approach may be termed “Active Matrix Electrowetting on Dielectric” (AM-EWOD). There are several advantages in using TFT based electronics to control an EWOD array, namely:                Driver circuits can be integrated onto the AM-EWOD array substrate. An example arrangement is shown in FIG. 7. Control of the EWOD array 42 is implemented by means on integrated row driver 76 and column driver 78 circuits. A serial interface 80 may also be provided to process a serial input data stream and write the required voltages to the array 42. The number of connecting wires 82 between the array substrate and external drive electronics, power supplies etc. can be made relatively few, even for large array sizes.        TFT-based electronics are well suited to the AM-EWOD application. They are cheap to produce so that relatively large substrate areas can be produced at relatively low cost.        It is possible to incorporate TFT-based sensing into Active Matrix controlled arrays. For example US20080085559 describes a TFT based active matrix bio-sensor utilising cantilever based arrays.        TFTs fabricated in standard processes can be designed to operate at much higher voltages than transistors fabricated in standard CMOS processes. This is significant since many EWOD technologies require EWOD actuation voltages in excess of 20V to be applied.        
In a number of EWOD applications it may be important to sense and/or control the temperature of droplets in an array. Examples of such instances may include:                Chemical or biochemical synthesis, where temperature control may be required to initiate and/or control chemical or biochemical reactions.        Detection of an endothermic or exothermic chemical reaction between droplets (i.e. the reaction absorbs or releases heat, and thus its detection indicates the occurrence of the reaction).        
An example of particular importance is the technique of polymerase chain reaction (PCR) for the amplification of DNA. PCR is a very well known technique, the details of which are well described in prior art, for example in chapter 1 of “The basics: PCR”, McPherson and Moller, Taylor and Francis (2nd Edition, 2006), ISBN 0-4153-5547-8. To implement PCR the DNA sample to be amplified must be mixed with various chemical reagents and then arranged to undergo a series of successive heating and cooling cycles. The total number of thermal cycles is typically 20-30. To perform PCR accurately and efficiently it is generally necessary to control the temperature of the chemical reagents reasonably accurately (typically to within a few degrees Celsius). In order to do this it is generally advantageous to implement some means of feedback control, an example of which is shown in FIG. 8. The chemical reagents 84 whose temperature is to be modified/controlled are arranged to be in thermal contact with a surface 86. A heater element 88 and temperature sensor element 90 are disposed upon this surface 86, in good thermal contact with the chemical reagents 84. The heater element 88 is electrically connected via 94 to a heater driver circuit 92 which is in turn connected via 102 to a processing unit 104, e.g. a CPU. The temperature sensor element 90 is connected via 96 to a temperature measurement circuit 98, which is in turn connected via 100 to the processing unit 104.
The whole arrangement operates such that the current supplied to the heater element 88 is determined in response to the temperature measured by the sensor element 90. The temperature can thus be more accurately controlled than in an arrangement only comprising a heater capability. A number of means for design of suitable feedback control circuits are known. For example WO2009/019658 (D. Fish et al., published Feb. 12, 2009) describes proportional integral (PI) and proportional integral differential (PID) schemes to optimise the thermal stability of the chemical reagent whose temperature is being controlled.
There exist a large number of well known techniques for sensing ambient temperature. Many of these methods can be found described in “Principles and methods of Temperature Measurement”, Thomas D. McGee, Wiley Interscience (1988). Page 230 of this publication describes a particularly convenient technique for measuring temperature using a forward biased PN junction (which can either be a diode or a diode connected transistor). The circuit symbol for the device is shown in FIG. 9. The device has two terminals, an anode 106 and a cathode 108. Forward bias operation corresponds to a positive potential V being applied to the anode 106 with respect to the cathode 108.
In forward bias operation the diode current I is given approximately by:
                    I        =                  S          ⁢                                          ⁢          W          ⁢                                          ⁢                      exp            ⁡                          (                              qV                mkT                            )                                                          (                  equation          ⁢                                          ⁢          1                )            
Where q is the electronic charge, k is Boltzmann's constant, T is the absolute temperature (in Kelvin), m is the diode ideality factor, a process dependent factor expressing the relative contributions of diffusion and recombination current, W is the diode width and S is a process dependent constant (which may also incorporate a temperature dependence). The relationship of equation (1) ceases to be valid when I exceeds a certain maximum limit (determined by the geometry of the device and processing details) because the current becomes limited by the diode's own self resistance.
It is easily shown that when such a device is forward biased (the anode voltage is positive with respect to the cathode voltage) with a constant current sink the voltage drop across the diode is proportional to the absolute temperature in Kelvin, specifically:
                    V        =                              mkT            q                    ⁢                      ln            ⁡                          (                              I                                  S                  ⁢                                                                          ⁢                  W                                            )                                                          (                  equation          ⁢                                          ⁢          2                )            
FIG. 10 shows a possible construction of a diode realised in a thin film process. The device is formed from a layer of semiconductor material consisting of a p+ doped region 112, an n-doped region 116 and an n+ doped region 114. Electrical connections, e.g. with metal are made to the p+ doped region 112 and n+ doped region 116 to form the anode 106 and cathode 108 terminals of the device. The diode p-n junction 118 is thus formed at the interface between the p+ doped region 112 and the n-doped region 116.
In operation in forward bias mode the device characteristics approximate to those described by equation 2, although typically the term S has a temperature dependence such that the total bias drop across the device at constant current is given approximately by:V=a−bT  (equation 3)
Where a and b depend on the current sunk and process dependent constants.
One possible circuit implementation for sinking a constant forward current through a diode is shown in FIG. 11.
A DC reference voltage Vref 126 is applied to the non-inverting terminal of the operational amplifier 124. The operational amplifier 124 works so as to maintain the same potential at the inverting terminal. Therefore the potential Vref is generated across the terminals of the resistor element 128 whose resistance is R. By consequence of Ohm's law, the current through this resistor element 128 is equal to I=Vref/R. Since (ideally) the input current of the operational amplifier is equal to zero, the current through the diode 110 is also equal to I. The anode of the diode 110 is coupled to the source or power supply voltage VDD 123. The circuit therefore sinks a constant current through the diode 110 and the voltage at the cathode is equal to VDD−V, where V is given by equation (2) above.
This voltage can then be converted into a digital signal by a suitable analogue-to-digital converter (ADC) 132. This ADC 132 may be of a standard type, for example consisting of a comparator followed by a counter. The DC voltage level at the ADC input is converted to a digitised output signal which represents the measured value of the temperature. The portion of the circuit termed the column output circuit 131, whose input is connected to the cathode of the diode 110, thus performs the function of sinking a constant current and measuring the potential at its input.
It will be apparent to one skilled in the art that there are many possible alternative implementations of a column output circuit 131 which may be devised to perform this function.
In practical implementations of the circuit of FIG. 11, depending on the value of the current being sunk through the diode 110, the required resistance value of the resistor element 128 may be quite large. In some instances this may be inconvenient to implement, for example due to both the physical layout size of the resistor. An additional disadvantage is that the resistivity of the material used to form the resistor element 128 may have undesirable temperature dependence. A well known technique for avoiding both of these problems is the use of a switched capacitor circuit to imitate a resistor element. FIG. 12 shows a simple implementation of such a switched capacitor resistor element. This arrangement is made to imitate a resistor element by applying a square wave pulse Φ to the gate of one of the transistors 134 and an anti-phase pulse Φbar to the gate of the other transistor 136. The technique is described in detail in “MOS Switched Capacitor Filters”, Brodersen et al, Proceedings of the IEEE, Vol. 67, num. 1 (Jan. 1979). The value of the effective resistance is determined from the frequency at which Φ and Φbar are switched and the value of the capacitor 135. One advantage of the switched capacitor implementation of a resistor element is that the layout area required to implement it may be less than that of a conventional resistor element. A further advantage is that the switched capacitor resistor element can be made relatively temperature independent. A further advantage is that it is possible to change the value of the resistance by changing the frequency of the pulses Φ and Φbar. Since the resistance can be changed by adjusting the frequency of the square wave pulses Φ and Φbar, it can be appreciated that the device can function as a variable resistor if some simple means of adjusting Φ and Φbar is available, e.g. a simple digital timing circuit.
A further disadvantage of the circuit of FIG. 11 is that the change in output voltage at the cathode of the diode 110 for a given change in the temperature may be quite small, for example only a few millivolts per degree Kelvin. This disadvantage is likely to be especially appreciable for the case where the temperature sensor is fabricated with thin film circuit components. The relatively poor quality and variability of the TFT components may make the design of an ADC capable of sensing an analogue voltage level to a few millivolts precision very difficult, and so the circuit of FIG. 11 may not be very sensitive. U.S. Pat. No. 3,791,217 (B. Stout et al.; issued Feb. 12, 1974) and JP2006073887 (T. Kuwabara et al.; published Mar. 16, 2006) describe a modified version of this implementation where the single diode element is replaced with n multiple diode elements connected in series. In this case the voltage drop across the series thin film diodes is given by the expression:V=n(a−bT)  (equation 4)
where n is the number of series connected diode elements. The change in output voltage for a given change in temperature has thus been increased by a factor n. The choice of value for n for a practical design of temperature sensor will depend on the detailed construction of the diode design and processing, the circuit implementation and operating voltages and the range over which the sensor must operate. For example, considering the circuit of FIG. 11, n should be made as large as possible whilst being careful to ensure that firstly the current sunk through the diodes is not so large that the self resistance of the diodes becomes important so that equation (5) ceases to be valid, and secondly that the voltage drop across the diodes as predicted by (5) does not exceed VDD−Vref, otherwise saturation at of the output of the sensor will occur.
A further disadvantage of the implementations of the implementation of FIG. 11 is that the output voltage is dependent on the process parameter S which may vary significantly from sample to sample. As a result accurate temperature measurement may require some form of initial calibration which may be costly from a manufacturing point of view. U.S. Pat. No. 3,430,077 (D. Bargen; issued Feb. 25, 1969) and U.S. Pat. No. 3,812,717 (G. Miller et al.; issued May 28, 1974) describe methods for overcoming this disadvantage by arranging for two measurements of the voltage drop across the diode to be made with different currents being sunk for each, I1 and I2 respectively. Assuming the diodes have the characteristic given by equation (1), for the first measurement the output voltage is given by
                              V          1                =                              nmkT            q                    ⁢                      ln            ⁡                          (                                                I                  1                                                  S                  ⁢                                                                          ⁢                  W                                            )                                                          (                  equation          ⁢                                          ⁢          5                )            
In the second case the output voltage is given by
                              V          2                =                              nmkT            q                    ⁢                      ln            ⁡                          (                                                I                  2                                                  S                  ⁢                                                                          ⁢                  W                                            )                                                          (                  equation          ⁢                                          ⁢          6                )            
Following measurement of the two voltage levels V1 and V2 these are then subtracted to give a result
                              V          s                =                                            V              1                        -                          V              2                                =                                    nmkT              q                        ⁢                          ln              ⁡                              (                                                      I                    1                                                        I                    2                                                  )                                                                        (                  equation          ⁢                                          ⁢          7                )            
The voltage difference Vs, depends on the ratio of the two currents and the process dependent parameter S has been eliminated.
It will be apparent to one skilled in the art that there are a number of possible methods for implementing this technique. For example the column output circuit can be modified to the arrangement 141 shown in FIG. 13. A digital subtraction circuit 133 has been added to the output of the circuit of FIG. 11, and the resistor element 128 replaced by a variable resistor element 140 implemented by means of the switched capacitor arrangement previously described. In this implementation two separate measurements of the voltage drop across the temperature sensor element (diode 110) can be made with a different value of current sunk through the temperature sensor element in each case. The current may be varied either by:                Changing the value of Vref 126. Accordingly the current sunk will be Vref1/R and Vref2/R respectively for the two values of Vref where R is the resistance of the resistor element 140        Changing the frequency of the pulses applied to the switched capacitor arrangement employed as the resistor element 140. Accordingly, the effective resistance can be made different for each of the two measurements, for example values R1 and R2. Thus currents equal to Vref/R1 and Vref/R2 would be sunk through the diode 110 for the first and second measurements respectively.        
In accordance with equation 7, the two voltage measurements obtained can be subtracted to give a result where the process dependent parameter S has been eliminated. This could be achieved by a number of standard techniques, for example the two voltages could be subtracted in the analogue domain by means of intermediate storage on reference capacitors, or they could be subtracted following conversion to digital signals by means of a simple digital subtraction circuit 133.
A similar technique for eliminating the measurement dependence on the parameter S is described in U.S. Pat. No. 5,829,879 (H. Sanchez et al.; issued Nov. 3, 1998). This is achieved by performing two measurements of the output voltage with different widths of diodes switched into the circuit.
For the first case the total diode width is W1 and the voltage drop across the diodes is given by:
                              V          1                =                              nmkT            q                    ⁢                      ln            ⁡                          (                              I                                  S                  ⁢                                                                          ⁢                                      W                    1                                                              )                                                          (                  equation          ⁢                                          ⁢          8                )            
In the second case the total diode width is W2 and the voltage drop is:
                              V          2                =                              nmkT            q                    ⁢                      ln            ⁡                          (                              I                                  S                  ⁢                                                                          ⁢                                      W                    2                                                              )                                                          (                  equation          ⁢                                          ⁢          9                )            
The difference between these voltages is then:
                              V          s                =                                            V              1                        -                          V              2                                =                                    nmkT              q                        ⁢                          ln              ⁡                              (                                                      W                    2                                                        W                    1                                                  )                                                                        (                  equation          ⁢                                          ⁢          10                )            
The dependence on the parameter S has again been eliminated and the voltage Vs depends on the ration of W1 and W2.
For many applications it is useful to be able to sense temperature separately at a number of spatial locations, i.e. in an array-based format. Whilst it may be possible to perform such a function by implementing multiple copies of a temperature sensor element and measurement circuit using methods already described, this is generally impractical if the array size is required to be large since a very large number of connections needs to be made to the substrate, or if the array element size is required to be physically small.
WO2009019658 describes an array-based temperature sensor integrated in a thin film process. The sensor element described could be a (temperature dependent) resistor element, a forward biased diode or a diode connected transistor. The array element circuit described is shown in FIG. 14. This consists of a temperature sensor element (diode) 142, a switch transistor 144, and a second switch transistor 146. The gate of each of the switch transistors is connected to a row select line RWS at 152, common to each array element within the same row. The temperature sensor element 142 is connected between the switch transistors as shown. Two column electrodes 148,150 are connected to each element 148 and 140. These connections are common to each element in the same column of the array.
External circuitry is provided to operate the array as shown in FIG. 15. The first column electrode 148 is connected to a power supply voltage VDD 156. The second column electrode 150 is connected to a column output circuit 141 for sinking a constant current and measuring the voltage developed. The column output circuit 141 may be comprised of a current source 158 and an output amplifier 160 using standard circuit techniques, for example as has already been described. The arrangement of the output circuit 141 is duplicated for each column of the array. In operation a voltage high level is applied to the RWS line for the row element being read out. The switch transistors 144 and 146 are thus both turned on. An electrical connection is therefore made between the power supply voltage VDD 156 and the anode of the temperature sensor element 142, and between the cathode of the temperature sensor element 142 and the column output circuit 141. The arrangement then operates in the same way as the single element temperature sensor already described. It will be apparent to one skilled in the art that the calibration techniques described for the single element temperature sensor can also be readily extrapolated to an array-based architecture, whereby each element in the array may be calibrated individually using one or more of the techniques already described.
It may be noted that in this the array implementation of WO2009019658, three separate voltage lines must be supplied to each array element, and two switch transistors are required within each array element. A disadvantage of this arrangement is that by having multiple switch elements and bias lines the manufacturing yield may be lower than if fewer devices and connections were used. A further disadvantage is that the physical layout footprint of the array element is also larger than if fewer devices and connections were used.
WO2009019658 further describes the variant pixel circuit 155 shown in FIG. 16, using two diodes 145 and 147 of different sizes. A first switching element transistor 146 connects the first diode 145 to a read line 148, and the second switching element 144 connects the second diode 147 to second column electrode line 150. The gates of the two transistors 144 and 146 are connected to the same row select line RWS. The cathodes of both the first diode 145 and the second diode 147 are coupled to ground GRD. The same current is passed through each diode 145,147 and the voltage difference between the anodes is proportional to temperature and can be measured by means of the arrangement already described.
Other arrangements for array based temperature sensing are also known. For example U.S. Pat. No. 6,633,656 (F. Picard; issued Oct. 14, 2003) describes an array of thermistors for fingerprint sensing. A temperature sensor array may be used for the detection or recognition of fingerprints according to the well-known means described. When a finger is placed in thermal contact with the temperature sensor array, individual elements within the array may register different temperature changes in accordance with whether they are in local proximity to a ridge or a valley of the fingertip structure. Thus by scanning multiple elements in an array thermal means can be used to register and image of the fingerprint.
WO2009019658 further describes array based circuitry for supplying a current to a heating element. The basic circuit element 504 is shown in FIG. 17. A switch element transistor 514 is addressed by means of a write line 510 and a row select line 512 connected at A1. When the switch is turned on by means of the row select line at A1, a voltage is programmed across capacitor 516. Depending on the value of the programmed voltage the transistor 508 may be turned on to some extent. This in turn results in a current flowing from VCC 506 to a ground line 502 through transistor 508 and a resistive heating element 518.